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07-05-2023, 05:15 PM
Trans-amp Part-2
In the attached PDF are six figures which show a progression from a concept to a working circuit using a single-ended output stage.
Fig-1: VE stage
We see a basic opamp gain stage configured as a "virtual earth" amplifier. IN+ is grounded and IN- receives Ein via Rin. Feedback is applied via Rfb which goes from IN- to the opamp output.
The opamp is ideal and has infinite input resistance at its input pins, so there will be no current in or out of these pins. The opamp has infinite gain, so the feedback loop will be accurate. In this case, Ein is connected to the left-hand end of Rin and the right-hand end of Rin is held at ground potential as the opamp maintains zero voltage difference between its input pins. The current through Rin has nowhere to go except through Rfb. The opamp makes its output be the correct polarity for the current to flow as it should.
Voltage gain is set by the feedback loop, as A = Rfb / Rin
Fig-2: Gm input
Here, we have replaced Rin with a transconductance element that will accept a voltage input and produce a current output. Let's say this ratio is 1mA out for 1V in, which is then a transconductance gm = 1mA/V
Rfb has a value of 220k, so if we apply 1V of input to the gm element, its current can only go through Rfb, which will generate a voltage at the output of the opamp of 1mA x 220k = 220V. This is an impressive opamp!
Fig-3: Basic SE tube stage
A standard single-ended (SE) tube power amp circuit is shown with typical values.
Va = 320V
Vs = 320V via 1k
Vk = 32V via Rk
Vg = 0 (ground via Rg=220k)
Rk = 510R
Ik = 63mA
Pa = 20W
The voltage gain of this circuit is typically 10 or less, and we can use Vk as a guide to the peak Ein required to achieve clipping. In this case, we will say Ein max is 32V.
Fig-4: SE tube stage as VE
Now we have added feedback elements around the SE tube stage.
Note that we are calling the signal at the tube plate driving the output transformer (OT) Eout.
Any tube stage that provides gain is inherently inverting, just as the VE stage inverts the signal. So, adding the Rin and Rfb is pretty straight-forward. With the opamp, the reference pin IN+ is at ground and then IN- is made to be at this same voltage. The tube has asymmetrical input pins, as it were, where the cathode is the reference and is sitting at +32V. Were we to somehow get the control grid g1 to be at the same voltage, the tube would pull uncontrolled current and melt itself down. To keep the universe stable, the g1 must be negative with respect to the cathode for the cathode current to be in a useful range. Ideally we want the same 32V difference between G1 and k as the stock circuit had.
We can add Rin quite easily since the tube grid is effectively at ground potential. Rfb ties to the Rin but we need a capacitor to block the 320V at the plate. This seems reasonable enough.
Now we encounter problems.
The raw gain of the stage is only 10 as we saw above and the feedback loop can only make the closed-loop gain LOWER than the open-loop gain. This tells us right away that where we might have already needed a hefty input signal of 32V peak, with feedback we will need even more. Say we make the gain A = 2. If Rfb is 220k as in Fig-2, then Rin would be 110k. Where the tube grid itself may not draw any current from Rin, the grid-leak Rg=220k definitely will. Rg introduces a serious complication to the circuit: it is absolutely needed to keep the tube biased properly by tying g1 to ground, but it forms a voltage divider with Rin and Rfb.
Because the tube gain is so low, the feedback loop has very little "power" inasmuch as we have hardly any gain to sacrifice to make the loop accurate.
The asymmetry of the input pins of the tube is also an issue. The control grid can be deemed as having infinite impedance, where the cathode has an impedance of the reciprocal of the tube's transconductance. In the case of a 6L6, gm=5mS or so, and the reciprocal is then about 200R. It would be ideal if the circuit could be given a lot of extra gain to make the feedback loop behave better.
Fig-5: SE with Gm input
We have added the same magical transconductance element that we used in Fig-2. We have moved the DC blocking cap to be between the virtual-earth node 'X' and the tube grid and its grid-leak resistor.
We will assume the Gm element can handle whatever voltage is at the end of Rfb, which should be 320V. The gm element is a current source that can be varied with a voltage input. Like all current sources, its output impedance is ideally infinite.
There is still the issue of Rg stealing away a bit of the input current and making the circuit inaccurate, but it is much better than for the previous figure.
The DC blocking cap is still within the feedback loop and cause phase issues that lead to instability. Fortunately, the OT is outside this loop and we do not have to contend with its primary and parasitic elements.
Fig-6: Practical but imperfect " Transie"
" Transie" is the name Menno van der Veen applied to the small SE version of the Trans Tube Amp.
In this form, the Gm element is shown as an n-channel jfet. This is done to emphasise the fact that it must be a depletion-mode fet type, but it is more likely to be a mosfet in a real circuit. The LND150 is a candidate here. Menno used a surface-mount type device, which suited his ultimate circuit realisation where an extensive input buffer and bias control circuit are fashioned as a module.
Rfb is directly connected to the tube grid, which looks a bit scary.
Rk has been increased in value so there can be voltage across QJN and R2, its source resistor. ideally, R2 has a voltage across it that is greater than peak Ein so that the QJN current never goes to zero. We see that the operating Vgk is still 32V as the original circuit had.
A safety diode is added so that during start-up the tube grid can never be too positive with respect to the cathode. Fortunately, as soon as Va appears, there will be current through R2, QJN, Rfb and the OT primary and QJN will, for the most part, keep 'X' at a reasonable voltage.
Rg is still shown but is entirely unnecessary. We could leave it in as a 1M or higher, if it placates our mind to do so.
The overall circuit performance is now quite transformed, as is usually the case with hybrid circuits of tubes plus solid-state elements. The output impedance at the tube plate is greatly reduced and the OT is driven very well and its distortions are reduced.The tube's distortion is also reduced and the circuit sensitivity with respect to Ein requirements has gone from tens of volts to ones of volts.
We still have the problem of Rk and Ck. Remember that k is an input terminal for the tube and the gain from k to A is the same as from g1 to A, meaning that the RC constant of Rk + Ck telescopes through the tube. There are a number of ways to reduce or eliminate this effect, each with its pros and cons.
Hifi hobbyists are enamored with using 3-terminal voltage regulators to bias tubes through their cathodes. The regulator is configured as a current source. This has a huge negative impact inasmuch as the regulator's frequency response is superimposed upon the circuit. This is not a good thing. The usual bandaid is to parallel the regulator with a massive-value cap, which is needed even if the regulator is made out of discrete components which is a much better implementation. Either approach requires dealing with potential instability while the tube warms up.
An alternative is to configure a fixed-voltage supply to replace Rk+Ck. Again, there must be a very wide bandwidth control and low output impedance for this to be useful, plus it requires another PT or winding on the PT.
A further alternative is to reduce Rk to 10R or less and use a bias control circuit to help control the circuit during turn-on and maintain the operating point during operation. This is how Menno tackled the issues, and his Trans Tube Amplifiers book explores the experiments and the results.
In the attached PDF are six figures which show a progression from a concept to a working circuit using a single-ended output stage.
Fig-1: VE stage
We see a basic opamp gain stage configured as a "virtual earth" amplifier. IN+ is grounded and IN- receives Ein via Rin. Feedback is applied via Rfb which goes from IN- to the opamp output.
The opamp is ideal and has infinite input resistance at its input pins, so there will be no current in or out of these pins. The opamp has infinite gain, so the feedback loop will be accurate. In this case, Ein is connected to the left-hand end of Rin and the right-hand end of Rin is held at ground potential as the opamp maintains zero voltage difference between its input pins. The current through Rin has nowhere to go except through Rfb. The opamp makes its output be the correct polarity for the current to flow as it should.
Voltage gain is set by the feedback loop, as A = Rfb / Rin
Fig-2: Gm input
Here, we have replaced Rin with a transconductance element that will accept a voltage input and produce a current output. Let's say this ratio is 1mA out for 1V in, which is then a transconductance gm = 1mA/V
Rfb has a value of 220k, so if we apply 1V of input to the gm element, its current can only go through Rfb, which will generate a voltage at the output of the opamp of 1mA x 220k = 220V. This is an impressive opamp!
Fig-3: Basic SE tube stage
A standard single-ended (SE) tube power amp circuit is shown with typical values.
Va = 320V
Vs = 320V via 1k
Vk = 32V via Rk
Vg = 0 (ground via Rg=220k)
Rk = 510R
Ik = 63mA
Pa = 20W
The voltage gain of this circuit is typically 10 or less, and we can use Vk as a guide to the peak Ein required to achieve clipping. In this case, we will say Ein max is 32V.
Fig-4: SE tube stage as VE
Now we have added feedback elements around the SE tube stage.
Note that we are calling the signal at the tube plate driving the output transformer (OT) Eout.
Any tube stage that provides gain is inherently inverting, just as the VE stage inverts the signal. So, adding the Rin and Rfb is pretty straight-forward. With the opamp, the reference pin IN+ is at ground and then IN- is made to be at this same voltage. The tube has asymmetrical input pins, as it were, where the cathode is the reference and is sitting at +32V. Were we to somehow get the control grid g1 to be at the same voltage, the tube would pull uncontrolled current and melt itself down. To keep the universe stable, the g1 must be negative with respect to the cathode for the cathode current to be in a useful range. Ideally we want the same 32V difference between G1 and k as the stock circuit had.
We can add Rin quite easily since the tube grid is effectively at ground potential. Rfb ties to the Rin but we need a capacitor to block the 320V at the plate. This seems reasonable enough.
Now we encounter problems.
The raw gain of the stage is only 10 as we saw above and the feedback loop can only make the closed-loop gain LOWER than the open-loop gain. This tells us right away that where we might have already needed a hefty input signal of 32V peak, with feedback we will need even more. Say we make the gain A = 2. If Rfb is 220k as in Fig-2, then Rin would be 110k. Where the tube grid itself may not draw any current from Rin, the grid-leak Rg=220k definitely will. Rg introduces a serious complication to the circuit: it is absolutely needed to keep the tube biased properly by tying g1 to ground, but it forms a voltage divider with Rin and Rfb.
Because the tube gain is so low, the feedback loop has very little "power" inasmuch as we have hardly any gain to sacrifice to make the loop accurate.
The asymmetry of the input pins of the tube is also an issue. The control grid can be deemed as having infinite impedance, where the cathode has an impedance of the reciprocal of the tube's transconductance. In the case of a 6L6, gm=5mS or so, and the reciprocal is then about 200R. It would be ideal if the circuit could be given a lot of extra gain to make the feedback loop behave better.
Fig-5: SE with Gm input
We have added the same magical transconductance element that we used in Fig-2. We have moved the DC blocking cap to be between the virtual-earth node 'X' and the tube grid and its grid-leak resistor.
We will assume the Gm element can handle whatever voltage is at the end of Rfb, which should be 320V. The gm element is a current source that can be varied with a voltage input. Like all current sources, its output impedance is ideally infinite.
There is still the issue of Rg stealing away a bit of the input current and making the circuit inaccurate, but it is much better than for the previous figure.
The DC blocking cap is still within the feedback loop and cause phase issues that lead to instability. Fortunately, the OT is outside this loop and we do not have to contend with its primary and parasitic elements.
Fig-6: Practical but imperfect " Transie"
" Transie" is the name Menno van der Veen applied to the small SE version of the Trans Tube Amp.
In this form, the Gm element is shown as an n-channel jfet. This is done to emphasise the fact that it must be a depletion-mode fet type, but it is more likely to be a mosfet in a real circuit. The LND150 is a candidate here. Menno used a surface-mount type device, which suited his ultimate circuit realisation where an extensive input buffer and bias control circuit are fashioned as a module.
Rfb is directly connected to the tube grid, which looks a bit scary.
Rk has been increased in value so there can be voltage across QJN and R2, its source resistor. ideally, R2 has a voltage across it that is greater than peak Ein so that the QJN current never goes to zero. We see that the operating Vgk is still 32V as the original circuit had.
A safety diode is added so that during start-up the tube grid can never be too positive with respect to the cathode. Fortunately, as soon as Va appears, there will be current through R2, QJN, Rfb and the OT primary and QJN will, for the most part, keep 'X' at a reasonable voltage.
Rg is still shown but is entirely unnecessary. We could leave it in as a 1M or higher, if it placates our mind to do so.
The overall circuit performance is now quite transformed, as is usually the case with hybrid circuits of tubes plus solid-state elements. The output impedance at the tube plate is greatly reduced and the OT is driven very well and its distortions are reduced.The tube's distortion is also reduced and the circuit sensitivity with respect to Ein requirements has gone from tens of volts to ones of volts.
We still have the problem of Rk and Ck. Remember that k is an input terminal for the tube and the gain from k to A is the same as from g1 to A, meaning that the RC constant of Rk + Ck telescopes through the tube. There are a number of ways to reduce or eliminate this effect, each with its pros and cons.
Hifi hobbyists are enamored with using 3-terminal voltage regulators to bias tubes through their cathodes. The regulator is configured as a current source. This has a huge negative impact inasmuch as the regulator's frequency response is superimposed upon the circuit. This is not a good thing. The usual bandaid is to parallel the regulator with a massive-value cap, which is needed even if the regulator is made out of discrete components which is a much better implementation. Either approach requires dealing with potential instability while the tube warms up.
An alternative is to configure a fixed-voltage supply to replace Rk+Ck. Again, there must be a very wide bandwidth control and low output impedance for this to be useful, plus it requires another PT or winding on the PT.
A further alternative is to reduce Rk to 10R or less and use a bias control circuit to help control the circuit during turn-on and maintain the operating point during operation. This is how Menno tackled the issues, and his Trans Tube Amplifiers book explores the experiments and the results.
